Mixer circuit, receiver comprising a mixer circuit, wireless communication comprising a receiver, method for generating an output signal by mixing an input signal with an oscillator signal

ABSTRACT

The invention relates to a mixer circuit comprising an input node for receiving an input signal, a first output node  202 , and a second output node  203 , voltage to current conversion means and switching means operatively coupled to each other and to the input node, the first output node and the second output node to generate a mixed input signal at the first output node and the second output node in response to an oscillator signal. In an embodiment the voltage to current conversion means comprises a first and a second voltage to current converter, implemented as N-MOSFETs M 2  and M 3,  with their gates connected to the input node. The drain of M 2  is connected to the first output node  202,  while the drain of the M 3  is connected to the second output node M 3.  The source of M 2  is connected to the switching node  221,  while the source of M 3  is connected to the second switching node  222.  The switches SW are arranged to couple the first switching node  221  to a first supply voltage VDD and the second switching node  222  to a second supply voltage VSS during a first phase of the oscillator signal, and the first switching node  221  to VDD and the second switching node  222  to VSS during a second phase of the oscillator signal. The mixer circuit according to the invention may operate at low supply voltages by using switches connected only to the supply voltages VSS and VDD. Mixing is achieved by voltage to current converters GM 1  and GM 2,  which are alternatingly activated by the switches SW.

The invention relates to a mixer circuit comprising an input node forreceiving an input signal, a first output node, and a second outputnode, voltage to current conversion means and switching meansoperatively coupled to each other and to the input node, the firstoutput node and the second output node to generate a mixed input signalat the first output node and the second output node in response to anoscillator signal.

The invention further relates to a receiver for receiving radiofrequency signals comprises an antenna section coupled to a receiversection, having a local oscillator for generating an oscillatorfrequency, being arranged to output a signal at another frequency.

The invention further relates to a wireless communication devicecomprises a receiver coupled to a signal processing section forprocessing the signal at the lower frequency generated by the receiver.

The invention further relates to a method for generating an outputsignal by mixing an input signal with an oscillator signal, whereby theoutput signal comprises a first output current and a second outputcurrent, in a mixer circuit comprising an input node for receiving theinput signal, a first output node for providing the first outputcurrent, and a second output node for providing the second outputcurrent, voltage to current conversion means and switching meansoperatively coupled to each other and to the input node, the firstoutput node, and the second output node to generate the output signal atthe first output node and the second output node in response to theoscillator signal.

A known mixer circuit as described in the opening paragraph comprises afirst N-channel type metal-oxide-semiconductor field effect transistor(N-MOSFET) for converting an radio frequency (RF) signal superimposed ona bias signal, applied at its gate, into a drain current at its drain,while its source is connected to a negative power supply rail. The knownmixer circuit further comprises a second and a third N-MOSFET with theirsources connected to the drain of the first N-MOSFET, the drain of thesecond N-MOSFET is a first current output node, and the drain of thethird N-MOSFET is a second current output node. The second and thirdN-MOSFET are operated as switches. During a first phase of a localoscillator signal the second N-MOSFET is conductive, thereby passing onthe drain current of the first N-MOSFET to the drain of the secondN-MOSFET as a first output current. During a second phase of the localoscillator signal the third N-MOSFET is conductive, thereby passing onthe drain current of the first N-MOSFET to the drain of the thirdN-MOSFET as a second output current.

Mixers are commonly used for frequency translation in radio frequency(RF) communications systems. The frequency translation results frommultiplication of the RF input signal with a “local oscillator” (LO)signal. In practice, mixers are preferably implemented using so-called“hard switching” via a large LO signal, which mathematically correspondsto multiplication with a square wave, instead of a sine wave. Thisrenders higher a conversion gain (2/π instead of ½), and a lower noiseFig. Especially in complementary metal-oxide-semiconductor (CMOS) andbipolar-metal-oxide-semiconductor (BiCMOS) integrated circuit processtechnologies, most mixer circuits exploit switching.

A known problem for the realization of analog circuits in current andfuture CMOS and BiCMOS technologies is a continuing trend in supplyvoltage reduction. This leads to non- or poorly conducting switches inthe so-called “middle voltage range”, which are somewhere between thepositive and the negative power supply voltages and substantiallydifferent from these supply voltages. This problem expresses itself inanalog and mixed analog-digital circuits comprising switches, such asanalog-to-digital converters, digital-to-analog converters, but also inthe known mixer circuit.

In operation the first N-MOSFET in the known mixer circuit for linearityof the voltage to current conversion it is required that there issufficient gate-source and drain-source voltage headroom to operate instrong inversion and saturation. On the other hand it is required thatthe gate-source voltages of the second and third N-MOSFET aresufficiently large to establish a low-ohmic current path between thedrain of the first N-MOSFET and the first and second current output nodeif the second, respectively the third N-MOSFET is switched on. This mayeither be achieved by operating the entire mixer circuit at asufficiently high power supply voltage or by operating the mixer circuitat a lower power supply voltage and driving the gates of the second andthird N-MOSFET separately at voltages well above the power supplyvoltage.

The first approach is disadvantageous, since it may require power supplyvoltages that are higher than allowed from an IC processing technologypoint of view, thereby deteriorating the reliability of the N-MOSFETsleading to a reduced operating lifetime of the mixer circuit. The secondapproach is disadvantageous, since the required driving circuitscomplicate circuit design, especially on high frequencies in theGiga-Hertz (GHz) range. Furthermore also in this approach reliabilitymay an issue, because the required drive voltages are above the powersupply voltage.

Amongst others it is an object of the invention to obtain a mixercircuit without requiring an increased power supply voltage or a drivingcircuit for supplying voltages outside the power supply voltage range.

To this end the invention provides a mixer circuit as defined in theopening paragraph which is characterized in that the voltage to currentconversion means comprises:

a first voltage to current converter having a first control electrodecoupled to the input node and a first main conductive path having afirst output electrode coupled to the first output node and a firstswitching electrode coupled to the switching means;

a second voltage to current converter having a second control electrodecoupled to the input node and a second main conductive path having asecond output electrode coupled to the second output node and a secondswitching electrode coupled to the switching means; and

the switching means is arranged to couple

the first switching electrode to a first supply voltage and the secondswitching electrode to a second supply voltage during a first phase ofthe oscillator signal, and

the first switching electrode to the second supply voltage and thesecond switching electrode to the first supply voltage during a secondphase of the oscillator signal.

During operation the switching nodes of the first and second switchablevoltage to current converters need to be drawn either to the first powersupply voltage or the second power supply voltage. This requiresswitching means for establishing low ohmic conductive paths in theproximity of either the first or the second supply voltage, instead ofswitching means for establishing low ohmic conductive path at a voltagein the middle voltage range. This circumvents the need for operating themixer circuit at a power supply voltage higher than desirable from areliability point of view. Neither is it necessary to include drivingcircuit for obtaining voltages outside the power supply voltage rangefor driving the switching means.

A receiver as described in the second paragraph is characterized in thatthe receiver section comprises a mixer circuit according to theinvention for mixing the oscillator signal with the radio frequencysignals.

A wireless communication device as described in the third paragraph ischaracterized in that the receiver is a receiver according to theinvention.

A method for generating an output signal by mixing an input signal withan oscillator signal as described in the fourth paragraph ischaracterized in that:

the voltage to current conversion means comprises:

a first voltage to current converter having a first control electrodecoupled to the input node and a first main conductive path having afirst output electrode coupled to the first output node and a firstswitching electrode coupled to the switching means;

a second voltage to current converter having a second control electrodecoupled to the input node and a second main conductive path having asecond output electrode coupled to the second output node and a secondswitching electrode coupled to the switching means; and

the switching means coupling

the first switching electrode to a first supply voltage and the secondswitching electrode to a second supply voltage during a first phase ofthe oscillator signal, and

the first switching electrode to the second supply voltage and thesecond switching electrode to the first supply voltage during a secondphase of the oscillator signal.

The above and other objects and advantageous features of the presentinvention will become more apparent from the following detaileddescription considered in connection with the accompanying drawings inwhich:

FIG. 1A shows a schematic diagram of a prior art mixer circuit;

FIG. 1B shows a schematic functional representation of the prior artmixer circuit of FIG. 1A;

FIG. 2A shows a schematic functional representation of a an embodimentof a mixer circuit according to the invention;

FIG. 2B shows a schematic functional representation of anotherembodiment of a mixer circuit according to the invention;

FIG. 2C shows a schematic functional representation of yet anotherembodiment of a mixer circuit according to the invention;

FIG. 3 shows graphs with simulated thermal output noise current densityversus a local oscillator frequency for both a mixer circuit accordingto the invention and a known mixer circuit;

FIG. 4 shows a schematic circuit diagram of a an embodiment of a mixercircuit according to the invention realized as an integrated circuit;

FIG. 5 shows graphs with measured conversion gain versus a localoscillator frequency for the mixer circuit according to the invention,shown in FIG. 4;

FIG. 6 shows graphs with measured linearity versus a local oscillatorfrequency for the mixer circuit according to the invention, shown inFIG. 4;

FIG. 7 shows graphs with measured output noise versus a local oscillatorfrequency for the mixer circuit according to the invention, shown inFIG. 4; and

FIG. 8 shows a schematic diagram of a receiver comprising a mixercircuit according to the invention.

In these Figs. identical parts are identified with identical references.

FIG. 1A shows a schematic diagram of a prior art mixer circuit. It is acommonly used active mixer. It consists of a first N-MOSFET M1, a secondN-MOSFET M2, a third N-MOSFET M3, and a load network LOAD. The source ofthe first N-MOSFET M1 is connected to the negative power supply railVSS, the is drain connected to an internal node N1, while an inputsignal VB+VRF is provided at the gate. The source of the second N-MOSFETis connected to internal node N1, the drain is connected to the loadnetwork LOAD, while a first local oscillator signal LO+ is provided atits gate. The source of the third N-MOSFET is connected to internal nodeN1, the drain is connected to the load network LOAD, while a secondlocal oscillator signal LO− is provided at its gate. Furthermore theload network LOAD is connected to the positive power supply rail VDD.The first N-MOSFET M1 forms a transconductance stage or voltage tocurrent converter. The second N-MOSFET M2 and the third N-MOSFET M3 areswitches.

The transconductance stage M1 is biased around a bias voltage VB and isdesigned to implement a linear voltage to current conversion of an inputvoltage signal VRF, superimposed on the bias voltage VB, into thevariation of the drain current of the first N-MOSFET M1. Forlinearization purposes a source degeneration resistor may be insertedbetween the source of N-MOSFET M1 and the negative power supply railVSS.

The switches M2 and M3 are driven by the first local oscillator signalLO+ and the second local oscillator signal LO−, which are in anti-phasewith respect to each other. Both local oscillator signals are balancedaround a bias voltage VBLO which is not indicated in FIG. 1A. To mimicthe multiplication with a square wave signal at the local oscillatorfrequency, the amplitude of both local oscillator signals LO− and LO+must be chosen sufficiently high to fully switch the transconductorcurrent, provided at the drain of the transconductance stage M1 toeither output current IOUT1, the drain current of N-MOSFET M1, or IOUT2,the drain current of N-MOSFET M2.

N-MOSFETs M2 and M3 are preferably operated in saturation to actalternatingly as cascode devices to N-MOSFET M1, thereby improving theoutput resistance and the linearity of the mixer circuit. Depending onthe application the load network LOAD may be different. For instance, itmay consist of two resistors respectively connecting the drains ofN-MOSFETs M2 and M3 to the positive power supply rail VDD. This providesa wideband voltage conversion gain. Alternatively the load network LOADmay be a tuned LC network to provide gain only in a narrow frequencyband. In either case the operation principle of the mixer circuit is thesame.

To provide in operation a good linearity the N-MOSFET M1 in mixercircuit shown in FIG. 1A must have sufficient gate-source anddrain-source voltage headroom: only if N-MOSFET M1 is well in stronginversion and saturation, the transconductance stage achieves a goodlinearity. To an IIP3 well above 0 dBm, typical minimum drain-sourcevoltage values for a 0.18 μm CMOS process are in the range of 0.5 voltor more. With threshold voltages around 0.5 volt, this means that theminimum voltage of the gates of N-MOSFETs M2 and M3, to switch thesedevices on, is typically higher than 1 volt. Moreover, a large overdrivevoltage for the switches M2 and M3 are required to achieve low switchresistances.

Therefore, either supply voltage well above 1 volt are required, or aswitch driver circuit, driving the gates of switches M2 and M3 wellabove VDD. Such drivers are not easily implemented at operatingfrequencies in the GHz range, especially when wide bandwidth is requiredand LC tanks are impractical. Moreover, the maximum allowed gate voltageis decreasing for new technologies due to required reliability of thegate oxide of MOSFETs.

To address these problems, folded topologies have been proposed, e.g.P-MOSFET switches following a N-MOSFET transconductance stage. Howeversuch a mixer circuit requires adding a bias current source that addssubstantial noise, unless significant voltage headroom is reserved (butthen the switch again becomes the problem). In other popular mixers,like the passive mixer, very similar problems occur, especially in downconversion mixers, where AC coupling often is not possible (e.g. zero IFarchitecture) or requires very large capacitors (low IF architecture).The essence of the problem is the same: achieving low switch resistanceat voltage levels in the middle range between the supply voltages isimpossible without driving gates outside the supply. This problembecomes even more severe in future processes with even thinnergate-oxides and lower supply voltages, while threshold voltage onlyscale down slowly. Alternative mixer architectures able to operate at alow supply voltage directly compatible with digital CMOS technology aretherefore desired.

FIG. 1B shows a schematic functional representation of the prior artmixer circuit of FIG. 1A. It is a simplified representation of the mixercircuit shown in FIG. 1A. The transconductance stage M1 is representedby a voltage to current converter, voltage controller current source, GMwith a first terminal connected to the internal node N1 and a secondterminal connected to the negative supply voltage VSS, generating acurrent I(V) under control of the input signal VB+VRF applied at acontrol node. The switches M2 and M3 are represented by a switch drivenby the logic signals LO, representing the local oscillator signal LO+,and its inverse {overscore (LO)}, representing the local oscillatorsignal LO−, switching the current generated by the voltage to currentconverter GM to a first output node OUT1 as a first output current IOUT1and a second output node OUT2 as a second output current IOUT2.

FIG. 2A shows a schematic functional representation of a an embodimentof a mixer circuit 200 according to the invention. The shown mixercircuit 200 is a so-called single balanced switched transconductancemixer. It comprises two matched transconductors or voltage to currentconverters GM1 and GM2. An input signal VB+VRF is applied to the controlterminals 201, 211 of both transconductors GM1 and GM2. An outputcurrent IOUT1 is provided at output terminal 202 of the transconductorGM1 and a switching terminal 203 is coupled to a first switching node221. An output current IOUT2 is provided at output terminal 212 of thetransconductor GM2 and a switching terminal 213 is coupled to a secondswitching node 222. By means of the switches SW the first switching node221 is switched to the negative supply voltage VSS during a first phaseLO of a local oscillator signal, while simultaneously the secondswitching node 222 is switched to the positive power supply VDD. Duringa second phase LO of the local oscillator signal the first switchingnode 221 is switched to the positive power supply VDD, wilesimultaneously the second switching node 222 is switched to the negativepower supply VSS.

The key to the mixer circuit according to the invention is theobservation that the problems related to the known mixer circuit asshown in FIG. 1 a and FIG. 1 b relate to requiring a conductive channelat a voltage level in the middle range between the supplies VSS and VDD.However, it is easily possible to make low ohmic switches, provided thattheir conductive channel is connected to VSS (N-MOSFET) or VDD(P-MOSFET). This can be relied upon even in future CMOS technologies,for the simple reason that digital logic circuits rely on thisfunctionality (inverters).

The mixer circuit 200 shown in FIG. 2A conceptually, illustrates how asingle balanced mixer circuit according to the invention can beconstructed using two matched transconductors GM1 and GM2 and switchesSW connected to supply voltages VSS and VDD only. The transconductorsGM1 and GM2 are alternatingly switched on by switching their respectiveswitching terminals 203, 213 to the negative supply voltage VSS, andswitched off by switching their respective switching terminals 203, 213,to the positive supply voltage VDD. As explained GM1 is on, if GM2 isoff, and the other way around. For matched transconductors and idealinstantaneous switching, either IOUT1 or IOUT2 is equal to the productGmVrf, just as in the known mixer circuit shown in FIG. 1A and FIG. 1B,whereby Gm represents the transconductance factor of the transconductorsGM1 and GM2 and VRF the input voltage signal. Actually, both said knownmixer circuit and the mixer circuit 200 implement the same mixerfunction in different ways: said known mixer by a voltage to currentconversion followed by current switching, the mixer circuit 200according to the invention by directly switching transconductors(activate either one of the two “Switched Transconductors” GM1 and GM2).

FIG. 2B shows a schematic functional representation of anotherembodiment of a mixer circuit 250 according to the invention. The shownmixer circuit 250 is a so-called double balanced switchedtransconductance mixer. It comprises four matched transconductors orvoltage to current converters GM1 a, GM1 b, GM2 a, GM2 b. An inputsignal RF+ is applied to the control terminals 251, 261 oftransconductors GM1 a and GM2 a respectively. An input signal RF− isapplied to the control terminals 254, 264 of transconductors GM1 b andGM2 b respectively. Output terminals 252, 265 of transconductor GM1 aand GM2 b respectively are coupled to a first output node 281 forproviding an output current IOUT1. Output terminals 262, 255 oftransconductors GM2 a and GM1 b respectively are coupled to a secondoutput node 282 for providing an output current IOUT2. Switchingterminals 253, 256 of transconductors GM1 a and GM1 b respectively arecoupled to a first switching node 271. Switching terminals 263, 266 oftransconductors GM2 a and GM2 b respectively are coupled to a secondswitching node 272. By means of the switch SW the first switching node271 is switched to the negative supply voltage VSS during a first phaseLO of a local oscillator signal, while simultaneously the secondswitching node 272 is switched to the positive power supply VDD. Duringa second phase {overscore (LO)} of the local oscillator signal the firstswitching node 271 is switched to the positive power supply VDD, wilesimultaneously the second switching node 272 is switched to the negativepower supply VSS.

FIG. 2C shows a schematic functional representation of yet anotherembodiment of a mixer circuit 290 according to the invention. As withthe mixer circuit 250 shown in FIG. 2B, mixer circuit 290 is a doublebalanced switched transconductance mixer. It comprises four matchedtransconductors or voltage to current converters GM1 a, GM1 b, GM2 a,and GM2 b respectively. An input signal RF+ is applied to the controlterminals 251, 261 of transconductors GM1 a and GM2 a respectively. Aninput signal RF− is applied to the control terminals 254, 264 oftransconductors GM1 b and GM2 b respectively. Output terminals 252, 265of transconductors GM1 a and GM2 b respectively are coupled to a firstoutput node 281 for providing an output current IOUT1. Output terminals262, 255 of transconductors GM2 a and GM1 b respectively are coupled toa second output node 282 for providing an output current IOUT2. Theswitching terminal 253 of transconductor GM1 a and the switchingterminal 263 of transconductor GM2 a are coupled to a first switch SW1.The switching terminal 256 of transconductor GM1 b and the switchingterminal 266 of transconductor GM2 b are coupled to a second switch SW2.By means of the switches SW1 and SW2 the switching terminals oftransconductors GM1 a and GM1 b are switched to the negative supplyvoltage VSS during a first phase LO of a local oscillator signal, whilesimultaneously the switching terminals of transconductors GM2 a and GM2b are switched to the positive power supply VDD. During a second phase{overscore (LO)} of the local oscillator signal the switching terminalsof transconductors GM1 a and GM1 b are switched to the positive powersupply VDD, while simultaneously the switching terminals oftransconductors GM2 a and GM2 b are switched to the negative powersupply VSS.

The single balanced mixer circuit 200 shown in FIG. 2A has a strongoutput signal at the LO-frequency, which can be cancelled in the doublebalanced mixer circuit 250. By adding the transconductors GM1 b and GM2b driven by an RF-signal RF− which is the anti phase version of theRF-signal RF+ driving the transconductors GM1 a and GM2 a, this isreadily implemented. The double balanced switched transconductor mixercircuit 250 has the same nominal conversion gain as the double balancedversion of the known mixer circuit shown in FIG. 1A.

Despite of the functional equivalence, there are also significantdifferences. Most notably, in the known mixer circuit of FIG. 1A thereis an internal node N1 between the transconductor GM and the outputnode, which renders bandwidth limitations due to parasitic capacitanceand also distortion and noise effects. This internal node is lacking inthe switched transconductor mixer circuits 200, 250 according to theinvention. Moreover, the switches SW to the negative supply voltage VSSconstitute a common-mode current path for the two active transconductorsto the output nodes 281, 282. This ideally renders a constantcommon-mode output current, for ideal instantaneous switching. Inpractice, switching transients occur with most energy concentrated at2f_(LO), whereby 2f_(LO) represents the frequency of the localoscillator signal. This can easily be filtered out by capacitors toground. These common-mode currents also come with noise, however thishardly harms the noise fig. of the mixer circuits 200, 250 according tothe invention.

As indicated above the noise current introduced by the switching devicesSW is a common mode noise current. Thus, this noise current cancels inthe differential output current, which is the difference IOUT1-IOUT2 ofthe first output current IOUT1 and the second output current IOUT2. Forthe known mixer circuit shown in FIG. 1A the situation is completelydifferent. This is because there is a direct noise current path betweenthe outputs OUT1 and OUT2: when the local oscillator signals LO+ and LO−have approximately the same value, both switch transistors M2 and M3conduct and have significant noise current, resulting in a noise peakaround the zero-crossing. Also, local oscillator noise is amplifiedduring this time interval. This noise comes on top of the noise of thetransconductance stage M1, and dominates at high frequencies where the“zero-crossing region” constitutes a large portion of the localoscillator signal cycle time period. A similar effect occurs in passivemixers. In contrast, the switched transconductor mixers 200, 250according to the invention do not show this effect, as noise generatedby the switches SW is common mode noise.

FIG. 3 shows graphs with simulated thermal output noise current densityversus a local oscillator frequency for both a mixer circuit accordingto the invention and a known mixer circuit. The vertical axis shows theoutput noise in pA/sqrt(Hz), while the horizontal axis shows the localoscillator frequency in GHz. Graph 301 shows the simulated thermaloutput noise in dependence upon the local oscillator frequency of aknown mixer circuit as shown in FIG. 1A. Graph 302 shows the simulatedthermal output noise in dependence upon the local oscillator frequencyof a mixer circuit according to the invention as shown in FIG. 2A.

In both cases the transconductor is implemented using N-MOSFETs withW/L=15/0.3, nominally biased at V_(GS)=V_(DS)=0.65 Volt (0.5 Voltthreshold voltage). The switches have W/L=15/0.18 (NMOST) and 30/0.18(PMOST), and are driven with a 0 dBm local oscillator power (50 ohmtermination, balanced signals around a common mode voltage V_(dd)/2).The conversion transconductance is around 1 mS, and the bandwidth ofboth mixers is around 4 GHz. FIG. 3 shows the simulated thermal outputnoise current density with a low-ohmic termination of both the switchedtransconductor mixer circuit according to the invention and the knownactive mixer circuit.

Clearly the output noise behavior for both mixer circuits is verydifferent: where the output noise of switched transconductor accordingto the invention, shown in graph 302, decreases (roughly following thefrequency roll-off of the conversion transconductance), the output noiseof the known mixer circuit, shown in graph 301, increases. The low noiseof the switched transconductor mixer circuit according to the inventionis highly desired, because Low Noise Amplifiers (LNAs) usually havedecreasing gain at high frequencies, thus increasing the relevance oflow mixer noise at high local oscillator frequencies.

FIG. 4 shows a schematic circuit diagram of a an embodiment of a mixercircuit 400 according to the invention realized as an integratedcircuit. Mixer circuit 400 is driven by a differential radio frequencyinput signal comprising a first component RF+ and a second componentRF−. The differential output signal of mixer circuit 400 comprises afirst component Vout1 and a second component Vout2. The differentialoscillator frequency driving the mixer circuit 400 comprises a firstcomponent LO+ and a second component LO+.

The mixer circuit 400 comprises four voltage to current converters ortransconductors, GM1 a, GM1 b, GM2 a, and GM2 b, implemented byN-MOSFETs M5, M6, M7, and M8 respectively. The first component of theinput signal RF+ is applied to the gates of M5 and M8, while the secondcomponent of the input signal RF− is applied to the gates of M6 and M7.The sources of M5 and M6 are connected to a first switching node 420,while the sources of M7 and M8 are connected to a second switching node421. The drains of M5 and M7 are connected to a first output node 410.The drains of M6 and M8 are connected to a second output node 411.

Note that the transistors M5-M8 require a bias voltage to be applied totheir gates for proper functioning. Therefore the first component of theinput signal RF+ comprises a DC bias component, Vbias, and asuperimposed AC component, +Vin, while the second component of the inputsignal RF− comprises the DC bias component, Vbias, and a superimposed ACcomponent, −Vin, which is an anti-phase version of the signal +Vin.

The switches comprises two N-MOSFETs M1 and M2 and two P-MOSFETs M3 andM4. The first component LO+ of the oscillator signal is applied to thegates of M1 and M3, while the second component LO− of the oscillatorsignal is applied to the gates of M2 and M4. The drains of M1 and M3 areconnected to the first switching node 420, while the drains of M2 and M4are connected to the second switching node 421. The sources of M3 and M4are connected to a positive switch power supply VDD,SW. The sources ofM1 and M2 are connected to a negative power supply VSS.

The mixer circuit 400 further comprises an active load circuit forconverting currents I1 and I2, into the first component VOUT1 and secondcomponent VOUT2 of the output signal. Currents I1 and I2 are thecombined drain currents of M5 and M7, and the combined drain currents ofM6 and M8 respectively. The load circuit comprises two P-MOSFETs M9 andM10. The sources of M9 and M10 are connected to a positive power supplyVDD, which may be the same as the switch power supply VDD,SW. The drainof M9 is connected to the first output node 410, while drain of the M10is connected to the second output node 411. The gates of M9 and M10 areconnected to an internal node 430. The gates of M9 and M10 are biased bymeans of a bias current source 431 with a first terminal connected tointernal node 430 and with a second node connected to a negative supplyvoltage, which may be the negative supply voltage VSS, providing a biascurrent IB. A first output resistor ROUT1 is connected between internalnode 430 and the first output node 410, while a second output resistorROUT2 is connected between internal node 430 and the second output node411. A first output capacitor COUT1 is connected between the firstoutput node 410 and a negative supply voltage, which may be VSS, while asecond output capacitor COUT2 is connected between the second outputnode 411 and a negative supply voltage, which may be VSS.

The mixer circuit 400 has been realized to verify the mixer circuitaccording to the invention experimentally. It is a down conversion mixerwas designed to operate at 1 Volt supply voltage. The mixer circuit 400,shown in FIG. 4 shows the schematic that was realized on chip: astraightforward simple implementation of the double balanced SwitchedTransconductor Mixer concept of FIG. 3. The transconductors are shown inthe dashed boxes (M5-M8). Somewhat arbitrarily, the transconductance waschosen around 1 mS. Transistors M1 and M2 implement the switches to VSS,while M3 and M4 implement the switches to VDD. They are driven byanti-phase sine-wave signals around a common voltage equal to theinverter switch threshold (close to VDD/2). Note that sine-waves areused here for experimental reasons, but full swing digital signals canalso be used, enhancing the compatibility with digital CMOS.

To generate an the differential output voltage signal VOUT1-VOUT2, ancurrent to voltage (I-V) converter must be added. This is implemented bythe common-mode current-absorption circuit with the two resistors, ROUT1and ROUT2, and two P-MOSFETs, M9 and M10, in the upper part of FIG. 4.However, as such this circuit has a rather low common-mode outputvoltage. By adding the bias current source 431 providing the biascurrent IB, the common-mode output voltage is shifted up to a valuearound 0.6 Volt, to fit in the 1 Volt supply voltage. The mixer circuit400 is designed for a maximum conversion gain of around 20 dB(ROUT1=ROUT2=10 kohm), which can be lowered adding an external resistorbetween the first and second output nodes 410 and 411. In that arediscussed in the following, this resistor was chosen in the middle ofthe gain range to achieve 12 dB conversion gain. The mixer wasfabricated in a standard industrial 0.18 μm CMOS process.

In measurements termination resistors of 50 ohm were added on chip forapplying the radio frequency input signals RF+ and RF− and theoscillator signals LO+ and LO−, for ease of measurement. The chip wasmeasured via wafer probing, using baluns for single to differentialconversion at the input. A differential probe was used to measure thedifferential output voltage, the difference of VOUT1 and VOUT2. The IFbandwidth was 2 MHz, limited by the input capacitance of the probe (>10MHz is easily obtainable with an on-chip load).

FIG. 5 shows graphs with measured conversion gain versus a localoscillator frequency for the mixer circuit according to the invention,shown in FIG. 4. The vertical axis shows the conversion gain in dB,while the horizontal axis shows the local oscillator (LO) frequency inGHz. Curves 501 and 502 represent the conversion gain of the mixercircuit 400 in dependence upon the local oscillator frequency.

The conversion gain as a function of frequency was measured using twobaluns with overlap in frequency range: one for the 300 MHz-3 GHz band,curve 501, and one for 2-18 GHz, curve 502. Despite of experimentalinaccuracies it can be concluded that the mixer has 12 dB conversiongain and around 4 GHz LO bandwidth, which is in reasonable agreementwith simulation. The current consumption of the mixer consists of a moreor less constant term of 180 μA for the transconductors core, and adynamic term determined by the switching (≈200 μA/GHz). Note that thepower consumption is low because the transconductance is rather low,resulting in a high equivalent input noise resistance. To achieve lessthan 15 dB noise fig. with respect to 50 ohm, roughly 10 times highertransconductance, i.e. 10 times more power consumption, is needed. At 1GHz this will result in roughly 4 mW power consumption.

FIG. 6 shows graphs with measured linearity versus a local oscillatorfrequency for the mixer circuit according to the invention, shown inFIG. 4. The vertical axis shows the 3^(rd) order input referredintercept point (IIP3) in dB, while the horizontal axis shows the localoscillator (LO) frequency in GHz. Curves 601 and 602 represent the IIP3of the mixer circuit 400 in dependence upon the local oscillatorfrequency. FIG. 6 shows the commonly used IIP3, since this describes thelinearity of the mixer circuit by means of one parameter, therebycircumventing the need to describe for instance both the output rangeand the distortion.

IIP3 as a function of frequency was measured using two baluns withoverlap in frequency range: one for the 300 MHz-3 GHz band, curve 601,and one for 2-18 GHz, curve 602. An IIP3 better than +4 dBm is typicallyachieved for 12 dB conversion gain. Simulations and experiments withvarying output resistor showed that this linearity is limited by theoutput swing. Actually conversion gain and IIP3 can be traded.Simulations showed that an IIP3 in excess of +10 dBm is possible if theoutput voltage swing is reduced.

FIG. 7 shows graphs with measured output noise versus a local oscillatorfrequency for the mixer circuit according to the invention, shown inFIG. 4. The vertical axis shows output noise in dB μV/sqrt(Hz), whilethe horizontal axis shows the local oscillator (LO) frequency in GHz.Curves 701 and 702 represent the output noise of the mixer circuit 400in dependence upon the local oscillator frequency.

The output noise as a function of frequency was measured using twobaluns with overlap in frequency range: one for the 300 MHz-3 GHz band,curve 701, and one for 2-18 GHz, curve 702. FIG. 7 shows the outputnoise as a function of frequency, measured at 1 MHz IF frequency. Thetrend is similar to the conversion gain fall off, in agreement with thesimulation results shown in FIG. 3. Also the values fit roughly to thenoise current expected from the transconductor core according tosimulation. The 1/f corner frequency was around 1 MHz.

Summarizing FIG. 4, FIG. 5, FIG. 6, and FIG. 7, a 1 Volt switchedtransconductor mixer has been realized in standard 0.18 μm CMOS, with0.5 Volt threshold devices. It can operate at such low supply voltage,compatible with future digital CMOS, because only switches with aconductive channel connected to either VSS or VDD are used. In contrastto traditional active and passive CMOS mixers, the noise produced by theswitch transistors is common-mode noise, which is rejected at thedifferential output. As a consequence, the output noise of the switchedtransconductor mixer does not increase with LO frequency, in contrast toknown mixer circuits.

It will be clear to a person skilled in the art that differentvariations on the mixer circuit of FIG. 4 are possible. For instance:

A complementary implementations with P-MOSFETs replaced by N-MOSFETs andthe other way around.

Using BJTs (bipolar junction transistors) instead of M5-M8 (preferredimplementation in BiCMOS process, as BJT have much less 1/f noise).

Adding degeneration resistors in series with the transconductors (M5-M8)to increase the linearity.

Combining the circuit of FIG. 3 with its complementary version, so thatbias currents of the N-type transconductors are re-used in the P-typetransconductors. In order to reduce the required minimum power supply inthis case, Vbias, of the N-type transconductance stage can be chosenequal to Vdd, while for the PMOSTs it can be chosen equal to Vss. DCbiasing can be done by (high) resistors to Vss and Vdd respectively,while capacitors can provide the signal coupling. The outputs of thecomplementary mixer halves can be DC coupled, which is important forzero-IF down-conversion mixer applications.

FIG. 8 shows a schematic diagram of a receiver comprising a mixercircuit according to the invention. The receiver 800 comprises anantenna section 801 and a receiver section 802. The antenna 801 isarranged for receiving radio frequency (RF) signals. The purpose of theantenna section 801 is to receive the RF signal and put it through tothe receiver section 802. It comprises an antenna 811 and it maycomprise a matching network 812 to match the impedance of the antenna811 to the input impedance of the receiver section 803. The receiversection comprises a mixer circuit 821 according to the invention. Thismay by for instance the mixer circuit shown in FIG. 4. The receiversection further comprises a local oscillator 822 for generating a localoscillator signal. The received RF signal is mixed with the localoscillator signal for generating an output signal 803 at an intermediatefrequency (IF). Note that FIG. 8 shows a basic concept. Usually areceiver comprises additional components, such as for instance alow-noise amplifier (LNA) to improve the performance.

A receiver as shown in FIG. 8 may be used in for instance a wirelesscommunication device, such as a handset of a mobile phone, although itwill be clear that other wireless communication devices are possibletoo. In such a wireless communication device the output signal 803 maybe further processed in a processing unit. The processing unit may forinstance to generate an audio signal in dependence upon the outputsignal 803.

In summary the mixer circuit according to the invention has a number ofattractive features.

For instance, it can operate at very low supply voltages compatible withdigital CMOS because all required switches connect nodes to either Vssor Vdd.

Further it is capable of high frequency operation because there is adirect connection of the transconductor output nodes to thecurrent-outputs Iout1 and Iout2, without an “internal” node like n1 inFIG. 1. In the mixer of FIG. 1, capacitance from node n1 to ground is animportant bandwidth limitation.

Moreover, the noise introduced by the switch transistors is COMMON MODEnoise for the two active transconductors and results in common modeoutput noise currents. This noise cancels in the differential outputcurrent IOUT1-IOUT2, which means that the switches render negligiblenoise contribution. For the mixer in FIG. 1 the situation is drasticallydifferent. The switches M2-M3 of the mixer circuit in FIG. 1 introducelarge peaks in the PSD of the noise at the zero crossings of theoscillator signal. This is because there is a direct noise current pathbetween both outputs. When the LO+ and LO− have approximately the samevoltage, both switches conduct and have significant noise current,resulting in noise peak around the zero crossing. This noise comes ontop of the noise of the transconductance stage, and dominates at highfrequencies where the “take-over region” constitutes a large part of thelocal oscillator signal period time. A similar effect occurs in passivemixers. In contrast, the switched transconductor mixer shows much lessnoise deterioration at high frequencies.

The switched transconductor mixer circuit according to the invention canwork with plain full swing digital oscillator signals. The common modevoltage and amplitude are less critical as in known active mixercircuits, where the switch transistor preferably should remain in stronginversion and saturation to work as cascodes for the transconductorstage. Also the fact that noise of the switches is less critical asmentioned above makes the LO generation easier.

Applications in which the mixer circuit according to the invention maybe used advantageously are for instance: low voltage CMOS transceiver instandard digital CMOS at low supply voltage; applications where highmixer bandwidth is required and the bandwidth of known active mixercircuits are limited by internal nodes; applications where a low noisefig. (NF) at a high local oscillator frequency is required and whereswitch take-over noise dominates noise fig.; and applications where useof a digital oscillator signal is desired. Digital oscillator signalgeneration becomes more and more feasible as speeds increase. Advantagesof digital oscillators are for instance their flexibility andprogrammability, and they profit from Moore's law with respect to thescaling of the physical dimensions of integrated circuits.

The embodiments of the present invention described herein are intendedto be taken in an illustrative and not a limiting sense. Variousmodifications may be made to these embodiments by persons skilled in theart without departing from the scope of the present invention as definedin the appended claims. It will be clear to a person skilled in the artthat different variations on the embodiments of the mixer circuitsaccording to the invention as shown in FIG. 2A, FIG. 2B, and FIG. 4 arepossible. For instance: in alternative, complementary implementations,P-MOSFETs may be replaced by N-MOSFETs and the other way around; bipolarjunction transistors (BJTs) may be used instead of M5-M8 (preferredimplementation in BiCMOS process, as BJT have much less 1/f noise); andeither of the mixer circuits of FIG. 2A and FIG. 2B may be combined withtheir complementary versions, so that bias currents of the N-typetransconductors are re-used in the P-type transconductors. In order toreduce the required minimum power supply in this case, the bias voltageof the N-type transconductance stage can be chosen equal to VDD, whilefor the P-MOSFETs it can be chosen equal to VSS. DC biasing can be doneby resistors, with a relatively high resistance, to VSS and VDDrespectively, while capacitors can provide the signal coupling. Theoutputs of the complementary mixer halves can be DC coupled, which isimportant for zero-IF down-conversion mixer applications.

Furthermore known types of load networks used for Gilbert mixers can beapplied, e.g.: resistors to VDD for wideband voltage conversion gain,while in addition capacitors can be added for low-pass filtering; atransimpedance amplifier; tuned band-pass LC networks for narrowbandapplications; a common mode current absorbing network with highdifferential resistance; and I/V converter via a transimpedanceamplifier (e.g. in down-conversion mixers, where the frequencies are lowenough to implement operational amplifiers (OPAMPs)).

1. A mixer circuit comprising an input node for receiving an inputsignal, a first output node, and a second output node, voltage tocurrent conversion means and switching means operatively coupled to eachother and to the input node, the first output node and the second outputnode to generate a mixed input signal at the first output node and thesecond output node in response to an oscillator signal, characterized inthat the voltage to current conversion means comprises a first voltageto current converter having a first control electrode coupled to theinput node and a first main conductive path having a first outputelectrode coupled to the first output node and a first switchingelectrode coupled to the switching means; a second voltage to currentconverter having a second control electrode coupled to the input nodeand a second main conductive path having a second output electrodecoupled to the second output node and a second switching electrodecoupled to the switching means; and the switching means is arranged tocouple: the first switching electrode to a first supply voltage and thesecond switching electrode to a second supply voltage during a firstphase of the oscillator signal, and the first switching electrode to thesecond supply voltage and the second switching electrode to the firstsupply voltage during a second phase of the oscillator signal.
 2. Amixer circuit as claimed in claim 1, characterized in that the firstvoltage to current converter and the second voltage to current converterare insulated gate field effect transistors whereby the gates are therespective control electrodes, the drains the respective outputelectrodes, and the sources the respective switching nodes.
 3. A mixercircuit as claimed in claim 1, characterized in that the switching meanscomprises: a first switch for coupling the first switching electrode tothe first supply voltage during the first phase of the oscillator signaland to the second supply voltage during the second phase of theoscillator signal; and a second switch for coupling the second switchingelectrode to the second supply voltage during the first phase of theoscillator signal and to the first supply voltage during the secondphase of the oscillator signal.
 4. A mixer circuit as claimed in claim3, characterized in that the first switch is a first inverter circuitwith a first switch output node coupled to the first switching node anda first switch input node for receiving the oscillator signal; and thesecond switch is a second inverter circuit with a second switch outputnode coupled to the second switching node and a switch second input nodefor receiving an inversed version of the oscillator signal.
 5. A mixercircuit as claimed in claim 4, characterized in that the first invertercircuit and the second inverter circuit comprise transistors
 6. A mixercircuit as claimed in claim 5, characterized in that: the first invertercircuit comprises a first switch insulated gate field effect transistorof a first type with its drain coupled to the first switch output node,its source coupled to the first supply voltage, and its gate coupled tothe first switch input node, and a second switch insulated gate fieldeffect transistor of a second type with its drain coupled to the firstswitch output node, its source coupled to the second supply voltage, andits gate coupled to the first switch input node; and the second invertercircuit comprises a third switch insulated gate field effect transistorof the first type with its drain coupled to the second switch outputnode, its source coupled to the first supply voltage, and its gatecoupled to the second switch input node, and a fourth switch insulatedgate field effect transistor of the second type with its drain coupledto the first switch output node, its source coupled to the second supplyvoltage, and its gate coupled to the second switch input node
 7. A mixercircuit as claimed in claim 1, characterized in that the mixer circuitcomprises a second input node for applying a second input voltage,whereby: the voltage to current conversion means comprises: a thirdvoltage to current converter with a third control electrode coupled tothe second input node, and a third main conductive path with a thirdoutput electrode coupled to the second output node and a third switchingelectrode coupled to the switching means; a fourth voltage to currentconverter with a fourth control electrode coupled to the second inputnode, and a fourth main conductive path with a fourth output electrodecoupled to the first output node and a fourth switching electrodecoupled to the switching means; and the switching means is arranged tocouple: the third switching electrode to the first supply voltage andthe fourth switching electrode to the second supply voltage during thefirst phase of the oscillator signal, the third switching electrode tothe second supply voltage and the fourth switching electrode to thefirst supply voltage during the second phase of the oscillator signal.8. A receiver for receiving radio frequency signals comprises an antennasection coupled to a receiver section, having a local oscillator forgenerating an oscillator frequency, being arranged to output a signal atanother frequency, characterized in that the receiver section comprisesa mixer circuit as claimed in claim 1 for mixing the oscillator signalwith the radio frequency signals.
 9. A wireless communication devicecomprises a receiver coupled to a signal processing section forprocessing the signal at the lower frequency generated by the receiver,characterized in that the receiver is a receiver as claimed in claim 8.10. A method for generating an output signal by mixing an input signalwith an oscillator signal, whereby the output signal comprises a firstoutput current and a second output current, in a mixer circuitcomprising an input node for receiving the input signal, a first outputnode for providing the first output current, and a second output nodefor providing the second output current, voltage to current conversionmeans and switching means operatively coupled to each other and to theinput node, the first output node, and the second output node togenerate the output signal at the first output node and the secondoutput node in response to the oscillator signal, characterized in thatthe voltage to current conversion means comprises a first voltage tocurrent converter having a first control electrode coupled to the inputnode and a first main conductive path having a first output electrodecoupled to the first output node and a first switching electrode coupledto the switching means; a second voltage to current converter having asecond control electrode coupled to the input node and a second mainconductive path having a second output electrode coupled to the secondoutput node and a second switching electrode coupled to the switchingmeans; and the switching means coupling: the first switching electrodeto a first supply voltage and the second switching electrode to a secondsupply voltage during a first phase of the oscillator signal, and thefirst switching electrode to the second supply voltage and the secondswitching electrode to the first supply voltage during a second phase ofthe oscillator signal.